Frequency synthesizer using digital pre-distortion and method

ABSTRACT

A method includes identifying a setting for a capacitor bank associated with a voltage-controlled oscillator in a closed-loop phase-locked-loop. The setting represents a combination of one or more capacitors in the capacitor bank. The method also includes estimating a gain introduced by the closed-loop phase-locked-loop when the oscillator operates using the identified setting. The method further includes estimating a response of a loop filter in the phase-locked-loop and identifying one or more coefficients for a digital filter using the identified gain and the identified loop filter response. The digital filter is operable to filter an input signal. In addition, the method includes modulating the filtered input signal using the phase-locked-loop to produce an output signal.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of prior U.S. patent application Ser.No. 10/606,682 filed on Jun. 26, 2003, now U.S. Pat. No. 6,833,767.

This application claims the benefit under 35 U.S.C. § 119(e) of U.S.Patent Application Ser. No. 60/458,515 filed on Mar. 28, 2003.

TECHNICAL FIELD

This disclosure is generally directed to communication systems and morespecifically to a frequency synthesizer using digital pre-distortion andmethod.

BACKGROUND

Wireless communication devices typically include a frequencysynthesizer, which is often a critical component of many techniques usedto modulate data onto an outgoing wireless signal. Conventionalmodulation techniques, such as Gaussian Frequency Shift Keyed (GFSK)modulation, are typically implemented using either I-Q up-conversion orby direct open-loop modulation of a voltage-controlled oscillator (VCO).The up-conversion approach is often susceptible to an undesired imagesignal and is typically costly in terms of analog area and power. Theopen-loop modulation approach is often prone to frequency drift in thevoltage-controlled oscillator and to gain errors.

SUMMARY

This disclosure provides a frequency synthesizer using digitalpre-distortion and method.

In one aspect, a method includes identifying a setting for a capacitorbank associated with a voltage-controlled oscillator in a closed-loopphase-locked-loop. The setting represents a combination of one or morecapacitors in the capacitor bank. The method also includes estimating again introduced by the closed-loop phase-locked-loop when the oscillatoroperates using the identified setting. The method further includesestimating a response of a loop filter in the phase-locked-loop andidentifying one or more coefficients for a digital filter using theidentified gain and the identified loop filter response. The digitalfilter is operable to filter an input signal. In addition, the methodincludes modulating the filtered input signal using thephase-locked-loop to produce an output signal.

One or more technical features may be present according to variousembodiments of this disclosure. Particular embodiments of thisdisclosure may exhibit none, some, or all of the following featuresdepending on the implementation. For example, in one embodiment, afrequency synthesizer using digital pre-distortion is provided. Inparticular, the frequency synthesizer uses closed-loop modulation of aphase-locked-loop to produce an output signal. The use of closed-loopmodulation may introduce various irregularities in the output signal. Asexamples, the output signal may suffer from loop gain in thephase-locked-loop and from variations in a loop filter used in thephase-locked-loop.

To at least partially compensate for the various irregularities in theoutput signal, the frequency synthesizer also includes a digitalpre-distortion filter. The pre-distortion filter distorts an inputsignal before it is modulated by the phase-locked-loop to produce theoutput signal. This distortion changes the input signal in such a way asto at least partially correct for the irregularities introduced by thephase-locked-loop. As a result, the output signal produced may havefewer or no irregularities. In a particular embodiment, thepre-distortion filter distorts the input signal such that the outputsignal is compliant with the Bluetooth standard.

This has outlined rather broadly several features of this disclosure sothat those skilled in the art may better understand the DETAILEDDESCRIPTION that follows. Additional features may be described later inthis document. Those skilled in the art should appreciate that they mayreadily use the concepts and the specific embodiments disclosed as abasis for modifying or designing other structures for carrying out thesame purposes of this disclosure. Those skilled in the art should alsorealize that such equivalent constructions do not depart from the spiritand scope of the invention in its broadest form.

Before undertaking the DETAILED DESCRIPTION below, it may beadvantageous to set forth definitions of certain words and phrases usedthroughout this patent document. The terms “include” and “comprise,” aswell as derivatives thereof, mean inclusion without limitation. Theterm. “or” is inclusive, meaning and/or. The phrases “associated with”and “associated therewith,” as well as derivatives thereof, may mean toinclude, be included within, interconnect with, contain, be containedwithin, connect to or with, couple to or with, be communicable with,cooperate with, interleave, juxtapose, be proximate to, be bound to orwith, have, have a property of, or the like. The term “controller” meansany device, system, or part thereof that controls at least oneoperation. A controller may be implemented in hardware, firmware, orsoftware, or a combination of at least two of the same. It should benoted that the functionality associated with any particular controllermay be centralized or distributed, whether locally or remotely.Definitions for certain words and phrases are provided throughout thispatent document, and those of ordinary skill in the art shouldunderstand that in many, if not most instances, such definitions applyto prior as well as future uses of such defined words and phrases.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of this disclosure and its features,reference is now made to the following description taken in conjunctionwith the accompanying drawings, in which:

FIG. 1 illustrates an example wireless device according to oneembodiment of this disclosure;

FIG. 2 illustrates an example frequency synthesizer according to oneembodiment of this disclosure;

FIG. 3 illustrates an example capacitor selection circuit according toone embodiment of this disclosure;

FIG. 4 illustrates an example gain calibration circuit according to oneembodiment of this disclosure; and

FIG. 5 illustrates an example method for digital pre-distortionaccording to one embodiment of this disclosure.

DETAILED DESCRIPTION

FIG. 1 illustrates an example wireless device 100 according to oneembodiment of this disclosure. The wireless device 100 illustrated inFIG. 1 is for illustration only. Other wireless devices could be usedwithout departing from the scope of this disclosure. Also, the wirelessdevice 100 in FIG. 1 has been simplified for ease of illustration andexplanation. Other or additional components could be included in thewireless device 100.

In the illustrated embodiment, the wireless device 100 includes anantenna 102. The antenna 102 facilitates the communication ofinformation over a wireless interface. The antenna 102 may represent anysuitable structure that is operable to facilitate the transmission orreception of wireless signals. As a particular example, the antenna 102may support the transmission and reception of radio frequency signals.

Radio frequency (RF) processing circuitry 103 is coupled to the antenna102. In this document, the term “couple” refers to any direct orindirect communication between two or more components, whether or notthose components are in physical contact with one another. The RFprocessing circuitry 103 processes the signals transmitted or receivedby the antenna 102. As particular examples, the RF processing circuitry103 could include one or more low-noise amplifiers, mixers, filters, andpower amplifiers.

A transceiver 104 is coupled to the RF processing circuitry 103. Thetransceiver 104 may receive an incoming signal received by the antenna102 and down-convert the signal to generate a baseband or intermediatefrequency signal. The transceiver 104 may also receive a baseband signalor an intermediate frequency signal and up-convert the signal fortransmission by the antenna 102. The transceiver 104 could include anyhardware, software, firmware, or combination thereof for facilitatingcommunication using the antenna 102. As a particular example, thetransceiver 104 could include a transmitter and a receiver.

Receive circuitry 106 is coupled to the transceiver 104. The receivecircuitry 106 receives and processes incoming signals received by thewireless device 100. For example, the receive circuitry 106 may receivethe baseband or intermediate frequency signal produced by thetransceiver 104 and process the signal to extract information containedin the signal. The receive circuitry 106 may include any hardware,software, firmware, or combination thereof for processing incomingsignals, such as a filter or decoder. In the illustrated example, theincoming signal represents voice information, and the extracted voiceinformation may be provided to a speaker 108 for presentation to a user.

Transmit circuitry 110 is coupled to the transceiver 104. The transmitcircuitry 110 receives and processes outgoing signals for transmissionby the wireless device 100. For example, the transmit circuitry 110 mayreceive voice information and process the information to produce abaseband or intermediate frequency signal. The baseband or intermediatefrequency signal may then be provided to the transceiver 104. Thetransmit circuitry 110 may include any hardware, software, firmware, orcombination thereof for processing outgoing signals, such as a filter oran encoder. In the illustrated example, the outgoing signal may includevoice information collected by a microphone 112.

A controller 114 is connected to the receive circuitry 106 and thetransmit circuitry 110. The controller 114 controls the operation andbehavior of the wireless device 100. For example, the controller 114could allow a user to mute the wireless device 100 so that no voiceinformation is transmitted by the wireless device 100. In this example,the controller 114 may instruct the transmit circuitry 110 to stoptransmitting information when the wireless device 100 is muted. Thecontroller 114 could represent any suitable controller, such as aprocessor. The logic executed by the controller 114 could be stored in amemory 116, which may represent any suitable storage and retrievaldevice or devices. In other embodiments, the controller 114 and thememory 116 could be omitted from the wireless device 100.

In one aspect of operation, the wireless device 100 may include afrequency synthesizer, such as a frequency synthesizer 118 in thetransceiver 104. The frequency synthesizer 118 uses closed-loopmodulation of a phase-locked-loop (PLL) to produce a phase or frequencymodulated output signal, such as the outgoing signal transmitted by theantenna 102. The use of closed-loop modulation may introduce variousirregularities in the output signal. To at least partially compensatefor the various irregularities in the output signal, the frequencysynthesizer 118 includes a digital pre-distortion filter, which distortsan input signal before it is modulated by the phase-locked-loop toproduce the output signal. This distortion changes the input signal in away that at least partially corrects for the irregularities introducedby the phase-locked-loop. As a result, the phase-locked-loop may producean output signal having fewer or no irregularities.

This frequency synthesizer 118 could be used in any suitable wirelessdevice 100. For example, the wireless device 100 could represent aBluetooth device, a Digital Enhanced Cordless Telephone (DECT) device,or a Global System for Mobile communication (GSM) device. In the exampleshown in FIG. 1, the wireless device 100 could represent a mobileheadset that may be worn by a user and that facilitates thecommunication of voice information to and from a mobile or othertelephone. The wireless device 100 could also represent a mobilehandset, a base station, or any other suitable device. The frequencysynthesizer 118 could also support any suitable modulation technique,including Frequency Shift Keyed (FSK) modulation, Gaussian FrequencyShift Keyed (GFSK) modulation, or Gaussian Minimum Shift Keyed (GMSK)modulation.

Although FIG. 1 illustrates one example of a wireless device 100,various changes may be made to FIG. 1. For example, other one-way ormulti-way communication devices could be used. As a particular example,FIG. 1 illustrates a wireless device 100 providing voice services. Otherdevices, such as devices connecting a computer to a printer or to apersonal digital assistant, could use the frequency synthesizer 118.Also, various components could be combined or omitted or additionalcomponents can be added to the wireless device 100 according toparticular needs. In addition, the transceiver 104 could be replaced bya transmitter, and the receive circuitry 106 could be omitted in thewireless device 100.

FIG. 2 illustrates an example frequency synthesizer 200 according to oneembodiment of this disclosure. The frequency synthesizer 200 illustratedin FIG. 2 may, for example, be used as the frequency synthesizer 118 inthe wireless device 100 of FIG. 1. The frequency synthesizer 200 shownin FIG. 2 is for illustration only. Other embodiments of the frequencysynthesizer 200 could be used without departing from the scope of thisdisclosure.

The frequency synthesizer 200 is operable to receive an input signal(f_(in)) 202 and produce an output signal (f_(out)) 204, where f_(in)represents a desired value for the instantaneous output frequency andf_(out) is the actual instantaneous output frequency. In the illustratedexample, the frequency synthesizer 200 includes a phase-locked-loop 206.The phase-locked-loop 206 is used to produce the output signal 204 bymodulating the input signal 202. In a particular embodiment, aBluetooth-compliant wireless device 100 can transmit or receive GFSKmodulated data on one of eighty 1 MHz channels spanning 2.402–2.480 GHz.In this embodiment, the frequency synthesizer 200 pre-distorts the inputsignal 202 before it is modulated, and the pre-distortion alters theinput signal 202 so the output signal 204 is compliant with theBluetooth standard.

In this example, the phase-locked-loop 206 includes a phasedetector/charge pump. (PD/CP) 208. The phase detector 208 identifies aphase difference between a reference signal (f_(ref)) 210 and a signalprovided by a frequency divider 216. The phase detector/charge pump 208outputs a signal proportional to this phase difference.

A loop filter 212 is coupled to the phase detector/charge pump 208. Theloop filter 212 filters the signal generated by the phasedetector/charge pump 208. The loop filter 212 may represent any suitablefilter, such as a low-pass filter or a band-pass filter. In otherembodiments, the loop filter 208 could be omitted from the frequencysynthesizer 200.

A voltage-controlled oscillator (VCO) 214 is coupled to the loop filter212. The oscillator 214 is operable to produce the output signal 204. Inone embodiment, the frequency of the signal produced by the oscillator214 is controlled by the signal produced by the phase detector/chargepump 208 and filtered by the loop filter 212. The oscillator 214 mayinclude any suitable oscillator operable to generate the output signal204.

In one embodiment, the oscillator 214 includes a capacitor bank that hasselectable capacitors. In this embodiment, the frequency of the outputsignal 204 generated by the oscillator 214 can be at least partiallycontrolled by selecting various combinations of capacitors in thecapacitor bank. In a particular embodiment, the capacitor bank in theoscillator 214 includes 64 different settings corresponding to a rangeof approximately 500 MHz with approximately 8 MHz per setting.

A frequency divider 216 is coupled to the oscillator 214 and to thephase detector/charge pump 208. The divider 216 is operable to alter thefrequency of the output signal 204 produced by the oscillator 214. Thedivider 216 then provides the altered output signal to the phasedetector/charge pump 208. By controlling how the divider 216 alters thefrequency of the output signal 204, the behavior of thephase-locked-loop 206 can be controlled.

A modulator 218 is coupled to the divider 216. The modulator 218 isoperable to receive and modulate the input signal 202. The resultingmodulated signal is provided to the divider 216, which uses themodulated signal to alter the output signal 204 generated by theoscillator 214. In this way, the modulator 218 controls the operation ofthe divider 216, which allows the modulator 218 to control the operationof the phase-locked-loop 206. The modulator 218 may represent anysuitable modulator, such as a Delta-Sigma (ΔΣ) modulator.

The frequency synthesizer 200 also includes a digital filter 220. Thedigital filter 220 performs data filtering on the input signal 202before the input signal is modulated by the phase-locked-loop 206. Thedigital filter 220 may represent any suitable digital filter. In aparticular embodiment, the digital filter 220 represents a digitalGaussian filter that performs filtering for the Bluetooth standard.

The phase-locked-loop 206 shown in FIG. 2 represents a closed-loopmodulator, which may introduce various irregularities into the outputsignal 204. For example, the phase-locked-loop 206 may introduce gaininto the output signal 204. Also, variations in the loop filter 212 mayalter the output signal 204 in undesired ways. Further, due to noiseconsiderations, it may be difficult to implement a closed-loopphase-locked-loop 206 with a wide enough loop bandwidth to modulate datadirectly. As a particular example, the phase-locked-loop 206 may have aloop bandwidth of 100 kHz and a data modulation bandwidth of greaterthan 100 kHz. If left uncompensated, the response of thisphase-locked-loop 206 could cause unwanted low-pass filtering of theinput signal 202.

To help compensate for these or other irregularities, a pre-distortionfilter 222 is provided in the frequency synthesizer 200. The digitalpre-distortion filter 222 is coupled to the digital filter 220 and themodulator 218. As described below, the digital pre-distortion filter 222is inserted in the data signal path to cancel the response of thephase-locked-loop 206. In effect, the digital pre-distortion filter 222alters the input signal 202 so that the variations produced by thephase-locked-loop 206 are at least partially reduced or eliminated. Thedigital pre-distortion filter 222 may represent any suitable digitalfilter. In one embodiment, the digital pre-distortion filter 222includes adjustable coefficients, which may be altered to change thebehavior of the digital pre-distortion filter 222.

As described below, various functions may be performed to control theoperation of the frequency synthesizer 200. This may include theexecution of a tuning algorithm that controls the selection ofcapacitors in the capacitor bank of the voltage-controlled oscillator214. This may also include an algorithm for estimating the loop gain ofthe phase-locked-loop 206 and an algorithm for estimating the responseof the loop filter 212. In addition, this may include an algorithm forcalculating the coefficient settings of the pre-distortion filter 222.In one embodiment, some or all of these algorithms could representon-chip algorithms. In one embodiment, the frequency synthesizer 200includes a controller 224 that executes the algorithms and a memory 226that stores the algorithms. Other embodiments of the frequencysynthesizer 200 could also be used.

In a particular embodiment, the frequency synthesizer 200 supports theBluetooth standard. The Bluetooth standard represents a frequencyhopping system, where receive-to-transmit turnaround time is 220 μs, andthe settling time of the frequency synthesizer 200 is less than thatwhen software and baseband overhead is considered. The variousalgorithms described above may be executed each time a frequency hop isperformed, which helps to account for:

(1) different capacitor select settings in the capacitor bank of theoscillator 214 over process, temperature, and the Bluetooth frequencyrange;

(2) different gains (and hence different loop gains) over process,temperature, and the Bluetooth frequency range; and

(3) different component values in the loop filter 212 over process andtemperature.

For many wireless communication systems, the time spent on onetransmission may be relatively short. Thus, for a given transmission,the temperature variation may be small, so the capacitor selection, gaincalibration, and loop filter calibration could be essentially static.

Although FIG. 2 illustrates one example of a frequency synthesizer 200,various changes may be made to FIG. 2. For example, the loop filter 212may be removed from the frequency synthesizer 200. Also, the digitalfilter 220 and the digital pre-distortion filter 222 could be combinedinto a single filter.

FIG. 3 illustrates an example capacitor selection circuit 300 accordingto one embodiment of this disclosure. The capacitor selection circuit300 may, for example, be used in the frequency synthesizer 200 of FIG. 2to select the capacitors used in the capacitor bank of thevoltage-controlled oscillator 214. Other embodiments of the capacitorselection circuit 300 may be used without departing from the scope ofthis disclosure.

In the illustrated example, the frequency synthesizer 200 is configuredas a digital frequency-locked loop. In this example, the frequency ofthe signal produced by the voltage-controlled oscillator 214 iscontrolled by selecting various combinations of capacitors in acapacitor bank. Each combination of capacitors may correspond to adifferent setting. In a particular embodiment, the capacitor bankincludes 64 different settings, which can be represented by a six-bitvalue.

The frequency synthesizer 200 may execute a capacitor selectionalgorithm to select a setting for the capacitor bank in the oscillator214. During the execution of this algorithm, the oscillator 214 receivesa common mode voltage (VCM) 302. In this embodiment, the input to thevoltage-controlled oscillator (VCO) 214 is forced to the common modevoltage 302, and the VCO 214 is in the center of its analog tuningrange. In other embodiments, a different voltage input to the VCO 214may be appropriate.

The signal produced by the oscillator 214 is supplied to a prescaler304. The prescaler 304 receives and divides the signal from theoscillator 214. For example, the prescaler 304 may divide the signalproduced by the oscillator 214 by eight. In effect, this decreases thefrequency of the signal produced by the oscillator 214 by a factor ofeight. The prescaler 304 may represent any suitable structure that candivide the frequency of a signal.

The prescalar 304 supplies the divided signal to a counter 306. Thecounter 306 is clocked by the prescaler 304 to produce an output countersignal. The counter 306 could represent any suitable counter, such as anM counter. The output signal from the counter 306 is supplied to are-sync unit 308. The re-sync unit 306 synchronizes the output signal ofthe counter 306 with the digital components of the capacitor selectioncircuit 300. The re-sync unit 308 may represent any suitable structurefor synchronizing a signal with a digital domain. In other embodiments,other circuits providing a digital estimate of the VCO's instantaneousoutput could be used.

A digital frequency discriminator 310 estimates the frequency of thesignal produced by the counter 306. In one embodiment, the digitalfrequency discriminator 310 receives the reference signal 210 and countsthe number of outputs produced by the counter 306 during each cycle ofthe reference signal 210. The digital frequency discriminator 310 thenoutputs an estimated frequency (f_(ESTIMATE)) 312 representing anestimate of the frequency of the counter 306. Because the output of thecounter 306 represents a scaled version of the signal output by theoscillator 214, the estimated frequency 312 also represents an estimateof the frequency of the oscillator 214. In a particular embodiment, thesignal 312 has a frequency of:

$f_{ESTIMATE} = \frac{\frac{f_{VCO}}{8}}{f_{REF}}$where f_(VCO) represents the frequency of the signal produced by theoscillator 214, and f_(REF) represents the reference frequency 210provided to the frequency synthesizer 200.

An adder 314 is coupled to the digital frequency discriminator 310. Theadder 314 is operable to receive the estimated frequency 312 produced bythe digital frequency discriminator 310 and a desired frequency(f_(DESIRED)) 316 identifying the desired frequency of the signal 312.The adder 314 then subtracts the estimated frequency 312 from thedesired frequency 316 to calculate a difference between the frequencies312, 316. The adder 314 may include any suitable structure operable toidentify the difference between two signals. An integrator 318 receivesthe results of the subtraction performed by the adder 314. Theintegrator 318 accumulates the differences produced by the adder 314.The integrator 318 may represent any suitable accumulator.

A low-pass filter 320 is coupled to the integrator 318 and filters theaccumulated differences produced by the integrator 318. For example, thelow-pass filter 320 may attenuate feedback variation caused by frequencyquantization noise. The filtered signals produced by the filter 320 areaveraged by an averaging unit 322. The resulting average 324 representsone of the capacitor settings in the capacitor bank of thevoltage-controlled oscillator 214.

In one aspect of operation, the voltage-controlled oscillator 214contains a bank of selectable capacitors, which allows the frequency ofthe voltage-controlled oscillator 214 to be coarsely set. A capacitorselection algorithm in the frequency synthesizer 200 is executed, whichconfigures the frequency synthesizer 200 as a digital frequency-lockedloop. For example, the capacitor selection algorithm could cause theoscillator 214 to begin receiving the common mode voltage 302. Thecapacitor selection algorithm then allows the capacitor selectioncircuit 300 to operate. The capacitor selection circuit 300 digitallyestimates the frequency of the oscillator 214 and chooses the setting ofthe capacitor bank based on this frequency estimate. The feedback pathin the capacitor selection circuit 300 may continuously adjust thecapacitor select setting of the oscillator 214 so that the estimatedfrequency 312 remains equal to or approximately equal to the desiredfrequency 316. The average capacitor select value 324 produced by theaveraging unit 322 may be stored for use during the remainder of atransmission by the frequency synthesizer 200. In one embodiment, thismechanism may be faster than conventional approaches, such assuccessive-approximation register (SAR) algorithms, because the outputof the oscillator 214 may be sampled more often. This may help to reducefrequency quantization error and reduce the time needed for an accuratefrequency estimate.

Although FIG. 3 illustrates one example of a capacitor selection circuit300, various changes may be made to FIG. 3. For example, other oradditional components could be used in the capacitor selection circuit300.

FIG. 4 illustrates an example gain calibration circuit 400 according toone embodiment of this disclosure. The gain calibration circuit 400 may,for example, be used in the frequency synthesizer 200 of FIG. 2 toidentify the gain introduced by the phase-locked-loop 206. Otherembodiments of the gain calibration circuit 400 may be used withoutdeparting from the scope of this disclosure.

In the illustrated example, the frequency synthesizer 200 is againconfigured as a digital frequency-locked loop. In this example, thefrequency of the signal produced by the voltage-controlled oscillator214 is controlled using an adjustable current digital-to-analogconverter (DAC) 402 in the charge pump 208 of the phase-locked-loop 206.

As shown in FIG. 4, the gain calibration circuit 400 includes thevoltage-controlled oscillator 214 of FIG. 2 and the components 304–322from the capacitor selection circuit 300 of FIG. 3. In this embodiment,the average capacitor select value 324 previously produced by thecapacitor selection circuit 300 is saved and supplied to thevoltage-controlled oscillator 214. A gain calibration algorithm is thenexecuted to estimate the open loop unity gain bandwidth ω_(LOOP) of thephase-locked-loop 206. This information is used to select a setting forthe DAC 402 in the charge pump 208. This information is also used tocalculate coefficients for the pre-distortion filter 222.

During execution of the gain calibration algorithm, the gain calibrationalgorithm configures the frequency synthesizer 200. For example, theloop filter 212 is configured so that a first capacitor 404 is shortedto the common mode voltage 302 (using a switch or other suitablemechanism). Also, in one embodiment, the current supplied to the chargepump 208 in the phase-locked-loop 206 is selectable by a three-bit DAC402. The phase detector 208 in the phase-locked-loop 206 includes UP andDOWN outputs, which are overridden so that that charge pump current iseither positive or negative. From the three-bit value for the chargepump 208 and by overriding the UP and DOWN outputs of the phase detector208, the charge pump 208 is configured as a four-bit, sign+magnitudecurrent-mode DAC.

At this point, the analog input voltage to the voltage-controlledoscillator 214 is set by the charge pump DAC current into a resistor 406in the loop filter 212, which controls the frequency of the oscillator214. The analog input voltage of the oscillator 214 may be determinedusing the formula:VCM+DAC _(SETTING) I _(LSB) R ₁where VCM is the common mode voltage 302 in the center of the analogtuning range of the oscillator 214, DAC_(SETTING) is the digital DACinput, I_(LSB) is the LSB size of the charge pump DAC, and R₁ is theloop filter resistor 406. On average, the output frequency of theoscillator 214 may be determined using the formula:f _(OUT) =f ₀ +DAC _(SETTING)(I _(LSB) R ₁ K _(VCO))where f₀ is the output frequency of the oscillator 214 with the analoginput voltage at VCM 302 for the saved capacitor select setting, andK_(VCO) represents the gain of the oscillator 214.

In one embodiment, the gain calibration algorithm performs the followingsteps:

(1) Set the desired frequency to f_(DESIRED)=f_(CHAN)−Δf, forcing theaverage output frequency to f_(CHAN)−Δf, where f_(CHAN) is the channelfrequency, and Δf is a frequency value chosen to make a good two-pointlinear approximation to the loop gain of the phase-locked-loop 206.

(2) After the loop settles, save the average charge pump DAC input valueas DAC_(NEG). Thus,f _(CHAN) −Δf=f ₀ −DAC _(NEG)(I _(LSB) R ₁ K _(VCO)).

(3) Set the desired frequency to f_(DESIRED)=f_(CHAN)+Δf, forcing theaverage output frequency to f_(CHAN)+Δf.

(4) After the loop settles, save the average charge pump DAC input valueas DAC_(pos). Thus,f _(CHAN) +Δf=f ₀ +DAC _(POS)(I _(LSB) R ₁ K _(VCO)).

(5) Calculate ΔDAC=DAC_(POS)−DAC_(NEG), where

${\Delta\;{DAC}} = {{{DAC}_{POS} - {DAC}_{NEG}} = {\frac{2\Delta\; f}{I_{LSB}R_{1}K_{VCO}}.}}$

(6) The approximate unity gain bandwidth of the phase-locked-loop may bedetermined using the formula:

${\omega_{LOOP} = \frac{I_{P}K_{VCO}R_{1}C_{1}}{2{\pi\left( {C_{1} + C_{2}} \right)}N}},$where I_(P) represents the current through the phase detector/chargepump 208, and N represents the scale by which the divider 216 dividesthe output signal 204.

(7) Using the ΔDAC and ω_(LOOP) values, given a desired loop gainω_(LOOP) _(—) _(DESIRED), a corresponding ideal DAC setting DAC_(IDEAL)can be determined as follows:

${DAC}_{IDEAL} = {\frac{\Delta\;{DAC}\;\omega_{LOOP\_ DESIRED}2{\pi\left( {C_{1} + C_{2}} \right)}N}{{2\Delta\;{fC}_{1}}\;}.}$

(8) In one embodiment, a 3-bit DAC 402 is used in the charge pump 208.As a result, the calculated DAC_(IDEAL) value for a desired loop gainmay not be precisely set. Therefore, DAC_(IDEAL) may be rounded down toDAC_(ACTUAL), which is the 3-bit DAC setting once the phase-locked-loop206 is configured as a standard phase-locked-loop 206. This allows anactual loop gain close to the desired loop gain.

(9) The error between the actual loop gain (based on the actual 3-bitcharge pump DAC) and an ideal charge pump current is calculated. Thiserror value is used later to calculate coefficients for thepre-distortion filter 222. Thus,

$\omega_{LOOP\_ ACTUAL} = {\frac{{DAC}_{ACTUAL}}{{DAC}_{IDEAL}}{\omega_{LOOP\_ IDEAL}.}}$

A resistor-capacitor (RC) tuner algorithm may also be executed by thefrequency synthesizer 200. In a particular embodiment, the RC tuneralgorithm is executed concurrently with the gain calibration algorithm.Also, in a particular embodiment, separate tuner algorithms could beexecuted for the metal capacitor 404 (which may represent a small loopfilter capacitor) and for a MOS capacitor 408 (which may represent alarge loop filter capacitor). Actual values for ω_(Z) (the loop filterzero) and ω_(P2) (the loop filter pole), ω_(Z) _(—) _(ACTUAL) and ω_(P2)_(—) _(ACTUAL), can be calculated and expressed as:ω_(Z) _(—) _(ACTUAL)=ω_(Z) _(—) _(NOMINAL)ω_(Z) _(—) _(ERROR)ω_(P2) _(—) _(ACTUAL)=ω_(P2) _(—) _(NOMINAL)ω_(P2) _(—) _(ERROR)where ω_(Z) _(—) _(NOMINAL) and ω_(P2) _(—) _(NOMINAL) represent theestimated loop filter zero and loop filter pole, and ω_(Z) _(—) _(ERROR)and ω_(P2) _(—) _(ERROR) represent the error calculated above.

After the capacitor select, gain calibration, and RC tuner algorithmshave been executed, the frequency synthesizer 200 may be configured as aFrac-N frequency synthesizer having a standard phase-locked-loop withthe charge pump DAC 402 set to DAC_(ACTUAL). The loop gain controlreduces settling time and noise variation.

In one embodiment, the transfer function from the input of theDelta-Sigma modulator 218 to the output of the phase-locked-loop 206 is:

${H_{PLL}(s)} \cong {\frac{f_{REF}\left( {1 + \frac{s}{\omega_{z}}} \right)}{1 + \frac{s}{\omega_{z}} + \frac{s^{2}}{\omega_{LOOP}\omega_{z}} + \frac{s^{3}}{\omega_{P2}\omega_{LOOP}\omega_{z}}}.}$

To compensate for this response of the phase-locked-loop 206, thefrequency response of the pre-distortion filter 222 may approximate theinverse of the phase-locked-loop's response. By using the transform s

f_(REF)(1−Z⁻¹), the transfer function of the desired pre-distortionfilter 222 may be determined using the formula:

${H_{PRE}(Z)} = \frac{1 + {B_{1}\left( {1 - Z^{- 1}} \right)} + {B_{2}\left( {1 - Z^{- 1}} \right)}^{2} + {B_{3}\left( {1 - Z^{- 1}} \right)}^{3}}{1 + {A_{1}\left( {1 - Z^{- 1}} \right)}}$where:

$B_{1} = {{\frac{f_{REF}}{\omega_{z}}B_{2}} = {{\frac{f_{REF}^{2}}{\omega_{LOOP}\omega_{z}}B_{3}} = {{\frac{f_{REF}^{3}}{\omega_{P2}\omega_{LOOP}\omega_{2}}A_{1}} = {\frac{f_{REF}}{\omega_{2}}.}}}}$Using the calculated values for ω_(LOOP) _(—) _(ACTUAL), ω_(Z) _(—)_(ACTUAL), and ω_(P2) _(—) _(ACTUAL), the coefficient values in thisformula may be calculated. These coefficient values may then be used inthe pre-distortion filter 222, which filters the input signal 202according to the coefficients. In one embodiment, the filtercoefficients may be updated in the pre-distortion filter 222 duringsettling of the phase-locked-loop 206 before data is transmitted, whichmay avoid transients in the digital filters 220, 222.

In other embodiments, compensation for the loop gain and the loop filtervariation could be done using a higher resolution charge-pump DAC 402and switches in the loop filter 212. These could be used to adjust thecomponent values of the loop filter 212.

In a particular embodiment, the frequency synthesizer 200 may beimplemented in 0.25 μm CMOS. In this embodiment, the frequencysynthesizer 200 could use approximately 35 mA and occupy approximately 4mm² die area. In this embodiment of the frequency synthesizer 200, thetime for the various algorithms to be executed and for the output signal204 to settle may take approximately 140 μs.

Although FIG. 4 illustrates one example of a gain calibration circuit400, various changes may be made to FIG. 4 For example, other oradditional components could be used in the gain calibration circuit 400.

FIG. 5 illustrates an example method 500 for digital pre-distortionaccording to one embodiment of this disclosure. The method 500 may bedescribed with respect to the frequency synthesizer 200 of FIG. 2. Themethod 500 could be used by any other suitable frequency synthesizer.

The frequency synthesizer 200 selects a capacitor setting for avoltage-controlled oscillator 214 at step 502. This may include, forexample, the controller 224 configuring the frequency synthesizer 200 asa frequency-locked loop. This may also include the controller 224allowing the capacitor select circuit 300 to operate and identify acapacitor setting that causes the estimated frequency 312 to equal orapproximately equal the desired frequency 316.

The frequency synthesizer 200 estimates the gain introduced by thephase-locked-loop 206 at step 504. This may include, for example, thecontroller 224 configuring the frequency synthesizer 200 as afrequency-locked loop. As particular examples, this may include thecontroller 224 shorting the capacitor 404 in the loop filter 212 to thecommon mode voltage 302 and overriding the UP and DOWN outputs of thephase detector 208. This may also include the controller 224 allowingthe gain calibration loop 300 to operate and identify the gain using theresults produced by the gain calibration circuit 400.

The frequency synthesizer 200 estimates the response of the loop filter212 at step 506. This may include, for example, the frequencysynthesizer 200 using the formulas above to identify values for ω_(Z)_(—) _(ACTUAL) and ω_(P2) _(—) _(ACTUAL).

The frequency synthesizer 200 and controller 224 identify coefficientsfor the pre-distortion filter 222 at step 508. This may include, forexample, the frequency synthesizer 200 identifying the transfer functionof the frequency synthesizer 200. This may also include the frequencysynthesizer 200 identifying the inverse of this transfer function toidentify another transfer function that is associated with the desiredpre-distortion filter 222. This may further include the frequencysynthesizer 200 identifying the coefficient values using the secondtransfer function. In addition, this may include the controller 224updating the pre-distortion filter 222 with the new coefficient values.

The frequency synthesizer 200 allows its output to settle at step 510.This may include, for example, the frequency synthesizer 200 waiting fora specified period of time. During this time, the pre-distortion filter222 may settle and begin operating with its new coefficient values. Thefrequency synthesizer 200 transmits data at step 512. This may include,for example, the frequency synthesizer 200 filtering an input signal 202using the new settings in the pre-distortion filter 222.

Although FIG. 5 illustrates one example of a method 500 for digitalpre-distortion compensation, various changes may be made to FIG. 5. Forexample, additional factors could be used to identify coefficient valuesfor the pre-distortion filter 222.

While this disclosure has described certain embodiments and generallyassociated methods, alterations and permutations of these embodimentsand methods will be apparent to those skilled in the art. Accordingly,the above description of example embodiments does not define orconstrain this disclosure. Other changes, substitutions, and alterationsare also possible without departing from the spirit and scope of thisdisclosure, as defined by the following claims.

1. A method, comprising: identifying a capacitance associated with avoltage-controlled oscillator in a phase-locked-loop; estimating a gainintroduced by the phase-locked-loop when the oscillator operates usingthe identified capacitance; estimating a response of a loop filter inthe phase-locked-loop; and identifying one or more coefficients for adigital filter using the identified gain and the identified loop filterresponse.
 2. The method of claim 1, further comprising: filtering aninput signal using the digital filter; and modulating the filtered inputsignal using the phase-locked-loop to produce an output signal.
 3. Amethod, comprising: identifying one or more distortions caused by aphase-locked-loop, the phase-locked-loop capable of modulating an inputsignal; pre-distorting the input signal based on the one or moreidentified distortions; and modulating the pre-distorted input signalusing the phase-locked-loop to produce an output signal, wherein thepre-distortion of the input signal at least partially reduces the one ormore identified distortions caused by the phase-locked-loop; whereinidentifying the one or more distortions comprises at least one of: (i)estimating a gain introduced by the phase-locked-loop when an oscillatorin the phase-locked-loop operates using a capacitance, and (ii)estimating a response of a loop filter in the phase-locked-loop based onat least one input value to a charge pump in the phase-locked-loop. 4.The method of claim 3, further comprising selecting the capacitanceassociated with the oscillator.
 5. The method of claim 3, whereinestimating the gain introduced by the phase-locked-loop comprises:setting a desired frequency associated with the phase-locked-loop to afirst value; identifying a first input value to a digital-to-analogconverter in the charge pump in the phase-locked-loop; setting thedesired frequency to a second value; identifying a second input value tothe digital-to-analog converter; identifying a difference between thefirst and second input values; and identifying the gain of thephase-locked-loop using the identified difference.
 6. The method ofclaim 5, wherein estimating the gain introduced by the phase-locked-loopfurther comprises: identifying an ideal input value for thedigital-to-analog converter using an estimated unity gain bandwidth ofthe phase-locked-loop and the identified difference; rounding down theideal input value to generate an actual input value; and identifying anactual gain introduced by the phase-locked-loop using the actual inputvalue.
 7. The method of claim 6, wherein estimating the response of theloop filter comprises: identifying an error between the ideal inputvalue for the digital-to-analog converter and the actual input value;and estimating the response of the loop filter using the identifiederror.
 8. The method of claim 3, wherein pre-distorting the input signalcomprises: setting one or more coefficients for a digital filter basedon the one or more identified distortions; and filtering the inputsignal using the digital filter, the digital filter capable ofdistorting the input signal prior to modulation.
 9. The method of claim8, wherein setting the one or more coefficients for the digital filtercomprises: identifying a transfer function from the input signal to theoutput signal; transforming the transfer function to generate atransformed transfer function; and calculating one or more values forthe one or more coefficients using the transformed transfer function.10. The method of claim 8, wherein filtering the input signal using thedigital filter comprises filtering the input signal using a firstdigital filter; and further comprising filtering the input signal usinga second digital filter prior to filtering the input signal using thefirst digital filter.
 11. The method of claim 3, wherein identifying theone or more distortions comprises both (i) estimating the gainintroduced by the phase-locked-loop, and (ii) estimating the response ofthe loop filter in the phase-locked-loop.
 12. An apparatus, comprising:a phase-locked-loop; and a controller capable of: identifying one ormore distortions caused by the phase-locked-loop; and pre-distorting aninput signal based on the one or more identified distortions; whereinthe phase-locked-loop is capable of modulating the pre-distorted inputsignal to produce an output signal, wherein the pre-distortion of theinput signal at least partially reduces the one or more identifieddistortions caused by the phase-locked-loop; and wherein the controlleris capable of identifying the one or more distortions by at least oneof: (i) estimating a gain introduced by the phase-locked-loop when anoscillator in the phase-locked-loop operates using a capacitance, and(ii) estimating a response of a loop filter in the phase-locked-loopbased on at least one input value to a charge pump in thephase-locked-loop.
 13. The apparatus of claim 12, wherein: thecontroller is further capable of selecting the capacitance associatedwith the oscillator in the phase-locked-loop.
 14. The apparatus of claim12, wherein the controller is capable of estimating the gain introducedby the phase-locked-loop by: setting a desired frequency associated withthe phase-locked-loop to a first value; identifying a first input valueto a digital-to-analog converter in the charge pump in thephase-locked-loop; setting the desired frequency to a second value;identifying a second input value to the digital-to-analog converter;identifying a difference between the first and second input values;identifying an estimated unity gain bandwidth of the phase-locked-loop;identifying an ideal input value for the digital-to-analog converterusing the estimated unity gain bandwidth and the identified difference;rounding down the ideal input value to generate an actual input value;and identifying the gain introduced by the phase-locked-loop using theactual input value.
 15. The apparatus of claim 14, wherein thecontroller is capable of estimating the response of the loop filter by:identifying an error between the ideal input value for thedigital-to-analog converter and the actual input value; and estimatingthe response of the loop filter using the identified error.
 16. Theapparatus of claim 12, further comprising a digital filter capable offiltering the input signal prior to modulation by the phase-locked-loop;and wherein the controller is capable of pre-distorting the input signalby setting one or more coefficients for the digital filter based on theone or more identified distortions.
 17. The apparatus of claim 16,wherein the controller is capable of setting the one or morecoefficients for the digital filter by: identifying a transfer functionfrom the input signal to the output signal; transforming the transferfunction to generate a transformed transfer function; and calculatingone or more values for the one or more coefficients using thetransformed transfer function.
 18. The apparatus of claim 16, wherein:the digital filter comprises a first digital filter; and furthercomprising a second digital filter capable of filtering the input signalprior to the first digital filter filtering the input signal.
 19. Theapparatus of claim 12, wherein: the controller and the phase-locked-loopform at least a portion of a transmitter; and further comprising anantenna capable of transmitting outgoing signals over a wirelessinterface, the transmitter capable of generating the outgoing signals.20. The apparatus of claim 12, wherein the controller is capable ofidentifying the one or more distortions by both (i) estimating the gainintroduced by the phase-locked-loop, and (ii) estimating the response ofthe loop filter.